{"id":2143,"date":"2026-08-16T15:52:11","date_gmt":"2026-08-16T07:52:11","guid":{"rendered":"https:\/\/www.trafopsu.com\/?p=2143"},"modified":"2026-08-16T15:52:11","modified_gmt":"2026-08-16T07:52:11","slug":"inductor-design-irms-isat-wire-diameter-srf","status":"publish","type":"post","link":"https:\/\/www.trafopsu.com\/ko\/inductor-design-irms-isat-wire-diameter-srf\/","title":{"rendered":"Inductor Design Guide: Irms, Isat, Wire Diameter, Core Material, SRF & Impedance"},"content":{"rendered":"

Inductor Design Guide: Irms, Isat, Wire Diameter, Core Material, SRF & Impedance<\/p>\n

\n

Choosing an inductor is never a single-number decision. The rated inductance and current on a datasheet are only the visible tip of a design that must balance heat, saturation, high-frequency loss, parasitic resonance and cost.<\/p>\n

On this page<\/strong><\/p>\n
    \n
  1. The six parameters at a glance<\/a><\/li>\n
  2. Irms \u2014 the heating limit, and wire diameter<\/a><\/li>\n
  3. Isat \u2014 the magnetic limit, set by core material<\/a><\/li>\n
  4. Wire diameter vs frequency \u2014 why 1 MHz needs Litz<\/a><\/li>\n
  5. Core material selection<\/a><\/li>\n
  6. Self-resonant frequency (SRF)<\/a><\/li>\n
  7. Impedance \u2014 what the circuit sees<\/a><\/li>\n
  8. How the parameters interlock<\/a><\/li>\n
  9. Worked example \u2014 4.2 mH \/ 25 A at 1 MHz\u00a0<\/a><\/li>\n
  10. Design checklist<\/a><\/li>\n
  11. Frequently asked questions<\/a><\/li>\n<\/ol>\n<\/div>\n

    In this guide we break down the six parameters engineers argue about most \u2014 Irms, Isat, wire diameter, core material, self-resonant frequency (SRF) and impedance<\/strong> \u2014 show how they constrain one another, and work through a demanding real-world example: a 4.2 mH \/ 25 A inductor operating at 1 MHz<\/strong> .<\/p>\n

    Whether you are specifying a power choke for a resonant converter, a filter inductor for an inverter, or a custom magnetic component, the same trade-offs apply. Read to the end for a practical design checklist and a FAQ section written for search visibility.<\/p>\n

    The Six Parameters at a Glance<\/h2>\n\n\n\n\n\n\n\n\n\n\n
    \ub9e4\uac1c\ubcc0\uc218<\/th>\nWhat it limits<\/th>\nDriven by<\/th>\n<\/tr>\n<\/thead>\n
    Irms<\/strong> (RMS current)<\/td>\nTemperature rise from I\u00b2R loss<\/td>\nWire cross-section, cooling, current density J<\/td>\n<\/tr>\n
    Isat<\/strong> (saturation current)<\/td>\nCore flux saturation (L drops)<\/td>\nCore material Bsat<\/sub>, core size, turns N<\/td>\n<\/tr>\n
    Wire diameter<\/strong><\/td>\nConductor area & skin effect<\/td>\nIrms, frequency f, skin depth \u03b4<\/td>\n<\/tr>\n
    Core material<\/strong><\/td>\n\u03bc, Bsat<\/sub>, losses, cost<\/td>\nFrequency, current, EMI, budget<\/td>\n<\/tr>\n
    SRF<\/strong> (self-resonant freq.)<\/td>\nParasitic turn capacitance<\/td>\nTurns, winding style, layout<\/td>\n<\/tr>\n
    Impedance Z<\/strong><\/td>\nHow the part behaves in circuit<\/td>\nL, f, DCR\/ACR, proximity to SRF<\/td>\n<\/tr>\n<\/tbody>\n<\/table>\n

    1. Irms \u2014 The Heating Limit, and How Wire Diameter Follows<\/h2>\n

    Irms is the continuous RMS current the inductor can carry while staying within its allowable temperature rise \u2014 typically 40\u00a0\u00b0C above ambient. It is fundamentally a thermal specification, not a magnetic one. The copper loss is:<\/p>\n

    P_cu = I_rms\u00b2 \u00d7 R_dc \u00d7 (1 + AC-loss factor)<\/div>\n

    To keep Irms high you need copper area. The required conductor cross-section comes straight from your chosen current density J (a design rule of thumb, not a physical constant):<\/p>\n

    A_cu (mm\u00b2) = I_rms \/ J\u00a0\u00a0\u00a0with\u00a0J \u2248 3\u20136 A\/mm\u00b2 for natural cooling,
    \nJ \u2248 6\u201310 A\/mm\u00b2 with forced air or potting<\/div>\n

    So wire diameter is not picked for looks \u2014 it is the direct consequence of Irms and your cooling budget. A 25 A inductor needs roughly 4\u20138 mm\u00b2 of total copper area. But at high frequency that area is only useful if the current can actually reach it (see skin effect below).<\/p>\n

    Practical note:<\/strong> Irms and Isat are independent limits. A part may saturate far below its thermal limit (small core, few turns) or overheat long before it saturates (thin wire, big core). Always check both.<\/div>\n

    2. Isat \u2014 The Magnetic Limit, Set by Core Material<\/h2>\n

    Isat is the DC (or low-frequency) current at which the inductance has fallen to a specified fraction of its initial value \u2014 usually 80% or 90%. Beyond this point the core can no longer support additional flux and the part loses its inductance. The peak flux density in the core is:<\/p>\n

    B_peak = (L \u00d7 I) \/ (N \u00d7 A_e)<\/div>\n

    At Isat, B_peak approaches the material’s saturation flux density Bsat<\/sub>. Two design levers follow:<\/p>\n

      \n
    • To survive a given current without saturating: raise N or Ae<\/sub>, or pick a core with more cross-sectional area \u2014 but more turns also raise DCR and lower SRF.<\/li>\n
    • To get more current headroom: a higher Bsat<\/sub> material (iron powder, sendust, MPP, high-flux) tolerates more current in the same volume than ferrite.<\/li>\n<\/ul>\n

      The stored energy tells the same story from the energy side. The energy an inductor must hold at its rated current is:<\/p>\n

      E = \u00bd \u00d7 L \u00d7 I_sat\u00b2<\/div>\n

      That energy must fit inside the core before it saturates, which is why a 4.2 mH \/ 25 A part is a large component: even at a conservative Isat of 30 A it must store about 1.9 J. Powder cores (distributed air gap) are favoured here because their soft, gradual saturation curve rides through current spikes without a sudden inductance collapse.<\/p>\n

      3. Wire Diameter vs Frequency \u2014 Why 1 MHz Demands Litz Wire<\/h2>\n

      At low frequency, wire diameter is set only by Irms and current density. At high frequency, the skin effect forces current into a thin surface shell. The skin depth in copper is:<\/p>\n

      \u03b4 = \u221a( \u03c1 \/ (\u03c0 \u00d7 f \u00d7 \u03bc\u2080) )\u00a0\u00a0\u2192\u00a0\u00a0\u03b4 \u2248 66 \u00b5m at 1 MHz<\/div>\n

      Rule of thumb: each individual conductor should have a diameter no larger than about 2\u03b4 (\u2248 0.13 mm at 1 MHz) or its centre stops carrying current and your effective copper area collapses. A solid 2.5 mm wire is therefore almost useless at 1 MHz \u2014 you would pay for copper that does nothing.<\/p>\n

      The standard fix is \ub9ac\uce20 \uc640\uc774\uc5b4<\/strong>: many individually insulated fine strands (each \u2264 2\u03b4) braided so that, on average, every strand occupies every radial position. For our 25 A example with J \u2248 4\u20135 A\/mm\u00b2 we need \u2248 5\u20136 mm\u00b2 of copper area, delivered as, for instance, hundreds of 0.071\u20130.10 mm strands grouped into 2\u20133 parallel Litz bundles. This keeps AC resistance close to DCR and is the only practical way to hit both 25 A and 1 MHz.<\/p>\n

      4. Core Material Selection \u2014 \u03bc, Bsat<\/sub>, Losses and Frequency<\/h2>\n

      The material choice sits at the centre of every trade-off. The usual suspects:<\/p>\n\n\n\n\n\n\n\n\n\n\n\n
      Material<\/th>\n\u03bcr<\/sub> (eff.)<\/th>\nBsat<\/sub><\/th>\nBest use<\/th>\n<\/tr>\n<\/thead>\n
      Ferrite (MnZn)<\/td>\n1500\u201315000<\/td>\n0.3\u20130.5 T<\/td>\n\u2264 ~500 kHz, high \u03bc, low cost<\/td>\n<\/tr>\n
      Ferrite (NiZn)<\/td>\n100\u20131500<\/td>\n0.3\u20130.4 T<\/td>\n0.5\u201310 MHz, low loss at RF<\/td>\n<\/tr>\n
      Sendust (Fe-Si-Al)<\/td>\n26\u2013125<\/td>\n\u2248 1.0 T<\/td>\nHigh-freq, soft saturation<\/td>\n<\/tr>\n
      Carbonyl iron powder<\/td>\n10\u2013100<\/td>\n\u2248 1.0\u20131.4 T<\/td>\n1 MHz+, distributed gap, low cost<\/td>\n<\/tr>\n
      MPP (Mo-Ni-Fe)<\/td>\n14\u2013550<\/td>\n0.7\u20130.8 T<\/td>\n1 MHz+, ultra-low loss, premium<\/td>\n<\/tr>\n
      High-Flux (Fe-Ni)<\/td>\n14\u2013160<\/td>\n\u2248 1.5 T<\/td>\nVery high Bsat<\/sub>, 25 A+ chokes<\/td>\n<\/tr>\n
      Nano-crystalline<\/td>\n15000\u2013100000<\/td>\n1.1\u20131.2 T<\/td>\n0.1\u20132 MHz planar power magnetics; very high \u03bc, very low core loss<\/td>\n<\/tr>\n<\/tbody>\n<\/table>\n

      For a 4.2 mH \/ 25 A \/ 1 MHz choke the priorities are: (a) enough effective permeability to reach 4.2 mH without an absurd turns count, (b) a high Bsat<\/sub> or soft-saturation curve to absorb 25 A, and (c) low core loss at 1 MHz. That points to a distributed-gap powder core \u2014 MPP or carbonyl iron for the lowest loss, high-flux or sendust where Bsat<\/sub> headroom matters most. Ferrite would need very high \u03bc to keep turns down but its low Bsat<\/sub> makes 25 A saturation risky; nano-crystalline is the standout choice for this 1 MHz spec<\/strong>: with very high permeability (\u03bcr<\/sub> in the tens of thousands) it reaches 4.2 mH with a fraction of the turns a powder core would need \u2014 the single biggest lever for keeping SRF above 1 MHz \u2014 while its ~1.2 T saturation and low high-frequency loss easily absorb 25 A. TrafoPSU’s O-26092 planar transformer is a real production example of 4.2 mH \/ 25 A at 1 MHz built on a nano-crystalline core. Distributed-gap powder cores (MPP, sendust, high-flux) remain a strong alternative where cost or soft-saturation behaviour is the priority.<\/p>\n

      5. Self-Resonant Frequency (SRF) \u2014 The Hidden Capacitor<\/h2>\n

      Every winding has parasitic capacitance between turns and layers. Together with the inductance it forms a parallel resonant tank. The self-resonant frequency is:<\/p>\n

      f_srf = 1 \/ ( 2\u03c0 \u221a(L \u00d7 C_p) )<\/div>\n

      Below f_srf the part is inductive; at f_srf its impedance peaks; above f_srf it behaves like a capacitor and is useless as an inductor. Two rules:<\/p>\n